Transconductance amplifier with nonlinear transconductance and low quiescent current

ABSTRACT

A composite transconductance amplifier is formed using a single transconductance amplifier with its output connected to a load via one or more resistors in series. The single transconductance amplifier has a linear transconductance (gm). As the current through the series resistors is increased, the voltage drops across the nodes of the resistors increase. Control terminals of separate drive circuits are connected to the various nodes and successively turn on as the current from the single transconductance amplifier slews more positive. Thus, the effective gm of the composite transconductance amplifier is based on the gm of the single transconductance amplifier and the currents contributed by the successively enabled drive circuits. Therefore, the gm is nonlinear. Pull-down drive circuits are also connected to the resistor nodes to successively pull down the current as the output from the single transconductance amplifier slews negative. The composite transconductance amplifier has low quiescent current.

FIELD OF THE INVENTION

This invention relates to transconductance amplifiers and, moreparticularly, to a transconductance amplifier having a selectablenonlinear transconductance (gm) while drawing a low quiescent current.

BACKGROUND

A transconductance amplifier receives an input voltage and outputs acurrent having some defined relationship to the input voltage. Thetransconductance (gm) of the amplifier is typically constant for a widerange of input voltages, but a nonlinear transconductance is desirablefor some applications. The transconductance is given as gm=ΔIout/ΔVinand may be variable for different ranges of ΔVin.

The output of a transconductance amplifier may be used to charge anddischarge a capacitor, control a MOSFET or bipolar transistor, or usedfor other purposes. Rapidly charging/discharging high capacitance loads,or driving large MOSFETs, may require a transconductance amplifier witha higher gm.

One example of the usefulness of a transconductance amplifier having anonlinear gm is in a feedback circuit such as used in a voltageregulator. The amplifier may receive a reference voltage and theregulator's divided output voltage. If the two voltages match (duringsteady state operation), the current output of the amplifier issubstantially zero and the output voltage of the regulator is notchanged. When the divided output voltage of the regulator becomesgreater than or less than the reference voltage, such as due to a changein the load, the non-zero output current from the amplifier causes theoutput voltage of the regulator to go down or up to again match thedivided output voltage to the reference voltage. Since there aretransient signals and delays involved in the operation of such afeedback circuit, stability is a concern. By the amplifier having a lowgm for small input voltage differences, the regulator has addedstability and, by the amplifier having a high gm for large input voltagedifferences, the regulator can quickly react to changing loads.

Such nonlinear gm transconductance amplifiers have many other uses.

The gm can be asymmetric, such as where a positive voltage differentialcauses the amplifier to have a particular gm and where a negativevoltage differential causes the amplifier to have a different gm. Suchan asymmetric gm may be where the amplifier is driving a device toemulate a diode.

FIG. 1 illustrates the current output of a transconductance amplifierhaving stepped gms and an asymmetric output. When there is a negativevoltage differential at the amplifier's input, the amplifier has a highgm, labeled H. The input may have an offset voltage, causing theamplifier to output substantially zero current during the offset range,labeled O. For a small positive voltage differential above the offset,the amplifier is configured to have a small gm, labeled S. For a largerpositive voltage differential, the amplifier is configured to have amedium gm, labeled M. For an even larger positive voltage differential,the amplifier is configured to have a high gm, labeled H.

Multiple transconductance amplifiers may be interconnected to achievethe desired characteristics of FIG. 1.

FIG. 2 illustrates one prior art technique for varying gms of acomposite transconductance amplifier, and FIG. 3 is a current vs.voltage curve obtained by the amplifier of FIG. 2.

The individual transconductance amplifiers are labeled as 10, 11, 12,13, and 14 and have respective transconductances of gm, gm1, gm2, gm3,and gm4. The gms may be the same or different. The amplifiers 10-14operate in different combinations to perform as a singletransconductance amplifier 16 having tailored characteristics. Differentvoltage offsets V1-V4 are shown coupled to the inverting inputs of theamplifiers. The polarities of the voltage offsets are identified. Thediodes D1-D4 are not part of the circuit but just convey the differentdirections of current output by the amplifiers 10-14. The compositeamplifier 16 is shown driving a capacitive load 20.

FIG. 3 shows the current vs. input voltage waveform for the amplifier 16of FIG. 2 as the input voltage Vin is swept from a negative voltage to apositive voltage. The waveform is symmetric about zero volts. The offsetvoltage levels are identified as V1-V4. The offset voltages determinewhen each amplifier 11-14 contributes to the output current. Theamplifier 10 is always contributing to the output current and operatesthrough the full voltage range. The amplifier 11 contributes currentwhen a positive input voltage differential exceeds V1, and its gm iscombined with that of the amplifier 10, shown as the medium gm M1 inFIG. 3. The amplifier 12 contributes current when the positive inputvoltage differential exceeds V2, and its gm is combined with that of theamplifiers 10 and 11, shown as the high gm H1. For a negative inputvoltage, the amplifier 13 contributes current when the negative inputvoltage differential exceeds V3, and its gm is combined with that of theamplifier 10, shown as the medium gm M2. The amplifier 14 contributescurrent when the negative input voltage differential exceeds V4, and itsgm is combined with that of the amplifiers 10 and 13, shown as the highgm H2. The composite amplifier 16 could have had an asymmetric output bychanging the gms of the individual amplifiers or the offsets.

Drawbacks with the design of FIG. 2 include the gms having discretelevels and each individual amplifier always drawing a quiescent current.More amplifiers can be added to increase the gm levels but this adds tothe quiescent current, complexity, and cost.

What is needed is an improved transconductance amplifier with anonlinear gm, where the quiescent current of the amplifier is much lowerthan that of the prior art design.

SUMMARY

In its simplest embodiment, a single transconductance amplifier with afixed gm is coupled to a load, such as a capacitive load, via aresistor. The current output by the amplifier creates a voltage dropacross the resistor. An NPN bipolar transistor has its base coupled toone end of the resistor and its emitter coupled to the other end of theresistor. Similarly, a PNP bipolar transistor has its base coupled toone end of the resistor and its emitter coupled to the other end of theresistor. The NPN transistor is progressively turned on by a positivecurrent from the amplifier as the voltage across the resistor exceedsthe positive base-emitter voltage necessary for turning on the NPNtransistor (e.g., 0.7 V). Similarly, the PNP transistor is progressivelyturned on by a negative current from the amplifier as the voltage acrossthe resistor exceeds the negative base-emitter voltage necessary forturning on the PNP transistor (e.g., −0.7 V). Thus, the gm of theamplifier is boosted for both positive and negative currents after anoffset that is determined by the value of the resistor. Therefore, thecircuit can be designed, or settable by the user, to have a small gmwithin any range of a zero voltage input, and a much larger gm outsideof that range. The gm can be asymmetric by the selection of NPN and PNPtransistors, or by using voltage offsets, or by using transistors inparallel. Further, since the currents supplied by the NPN and PNPtransistors smoothly ramp up after the voltage drop across the resistorreaches the associated thresholds, the gm variation is continuous andnot discrete. Therefore, the quiescent current is just that of a singletransconductance amplifier at the lowest gm, where the circuit mayoperate at for a majority of the time. The range of gms may be made veryhigh with no added quiescent current required.

In a more complex embodiment, multiple resistors are connected in seriesbetween the transconductance amplifier and the load to create a resistordivider having taps. Various drive circuits are connected to thedifferent taps so as to be enabled at different current level outputs ofthe transconductance amplifier. Each drive circuit may produce adifferent current to make the combined output current into the loadexponential or any other non-linear function of the input voltage. Thedrive circuits draw no current until enabled.

If the transconductance amplifier is used in a feedback loop, such as ina regulator which matches its output voltage to a reference, it isrelatively easy to compensate (for improving stability) due to its lowgm when the inputs are balanced (in steady state). The transconductanceamplifier has a high gm only when slewing.

Such a transconductance amplifier is also particularly suited fordriving high capacitance loads, including large MOSFETs, and forallowing customization of the gm for different capacitive loads. Otheruses include driving devices to emulate diodes.

Various other embodiments are described.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a possible current vs. voltage waveform of a transconductanceamplifier having an asymmetric nonlinear transconductance (gm).

FIG. 2 illustrates one example of a prior art composite transconductanceamplifier composed of a plurality of individual transconductanceamplifiers and voltage source offsets.

FIG. 3 is a possible current vs. voltage waveform of the nonlineartransconductance amplifier of FIG. 2, illustrating a symmetricalnonlinear gm.

FIG. 4 illustrates one embodiment of the invention, where a singletransconductance amplifier is used to provide a symmetric or asymmetricnonlinear gm, so quiescent current is low.

FIG. 5 illustrates another embodiment of the invention, where a singletransconductance amplifier is used to provide a symmetric or asymmetricnonlinear gm, so quiescent current is low, where the circuit provides alow gm at low differential voltages and a much higher gm outside of thatrange.

FIG. 6 illustrates another embodiment of the invention, where a singletransconductance amplifier is used to provide a symmetric or asymmetricnonlinear gm, so quiescent current is low, where tapped voltagesselectively enable drive circuits depending on the current output of thetransconductance amplifier.

FIG. 7 is a more detailed schematic of the transconductance amplifier ofFIG. 6.

FIG. 8 is a more detailed schematic of a selectable pull-up drivecircuit in the circuit of FIG. 6.

FIG. 9 is a more detailed schematic of a selectable pull-down drivecircuit in the circuit of FIG. 6.

Elements that are the same or equivalent are labeled with the samenumeral.

DETAILED DESCRIPTION

FIG. 4 illustrates one embodiment of the invention. The transconductanceamplifier 30 may be any conventional transconductance amplifier that hasa fixed transconductance (gm) within a certain range of input voltages.The output current I1 of the amplifier 30 is controlled by the inputvoltage Vin applied to its inverting and non-inverting input terminals.This current I1 generates a voltage drop across a resistor 32 in seriesbetween the amplifier 30 and a load 34, which is represented by a loadcapacitance, but may be any type of load. The load 34 may be a capacitorin a voltage regulator feedback network, a MOSFET, or any other type ofload where a nonlinear gm may be beneficial.

A bipolar NPN transistor 36 has its base coupled to one end of theresistor 32 and its emitter coupled to the other end of the resistor 32.Similarly, a bipolar PNP transistor 38 has its base coupled to one endof the resistor 32 and its emitter coupled to the other end of theresistor 32. The base currents are assumed to be negligible in theexample.

Assuming the current is a positive current (Vin is positive), the NPNtransistor 36 will not turn on until the voltage drop across theresistor 32 exceeds the base-emitter turn-on voltage (Vbe) of the NPNtransistor 36, which may be about 0.7V. Other types of transistors maybe used. Therefore, for low currents output by the amplifier 30, the NPNtransistor 36 does not add current to the load 34 so the overall gm ofthe circuit is relatively low (that of the transconductance amplifier30). As the current I1 is increased due to an increase in Vin, the NPNtransistor 36 turns on and contributes current into the load 34. Thecombined currents are shown as I2. The value of the resistor 32 isselected to determine the I1 threshold current needed to begin turningon the NPN transistor 36. Since the pull-up current provided by the NPNtransistor 36 smoothly increases as the current I1 exceeds the thresholdcurrent, there will be no discontinuities in the gm.

Similarly, the PNP transistor 38 is progressively turned on by anegative current from the amplifier 30 as the voltage across theresistor 32 exceeds the negative base-emitter voltage necessary forturning on the PNP transistor 38 (e.g., −0.7 V). The PNP transistor 38then contributes pull-down current to the negative current I1.

Thus, the gm of the amplifier is smoothly boosted for both positive andnegative currents after an offset that is determined by the value of theresistor. Therefore, the circuit can be designed, or settable by theuser, to have a small gm within any range of a zero differential voltageinput and a much larger gm outside of that range. The increased gm isexponential.

Since only a single transconductance amplifier 30 is used, the quiescentcurrent for the circuit of FIG. 4 is equal to the quiescent current of asingle transconductance amplifier, in contrast to the quiescent currentof the circuit of FIG. 2. The range of gms may be made very high with noadded quiescent current required and no further circuitry needed.

If the circuit of FIG. 4 was used in a feedback network, the feedbackloop would not be affected by the bipolar transistors during steadystate conditions, so the gm of the transconductance amplifier would beeasy to compensate to achieve the desired stability.

If high gms were desired, requiring added base current into the bipolartransistors, a multi-stage Darlington circuit can be used to boost thebase current. If MOSFETs were used instead of bipolar transistors, theneed for base drive is eliminated, but the gate capacitance may limitthe slew capability of the system.

FIG. 5 illustrates an added feature of a selectable offset for the NPNtransistor 36 and the PNP transistor 38 so that the transistors can beturned on at different current levels to achieve a tailored asymmetry ofthe I vs. V waveform. For example, the voltage source offset of Vo1 maysubtract or add voltage to the base of the NPN transistor 36 to cause itto turn on at a selected current I1 output from the transconductanceamplifier 30. Similarly, the voltage source offset of Vo2 may subtractor add voltage to the base of the PNP transistor 38 to cause it to turnon at a selected current I1 output from the transconductance amplifier30.

FIG. 6 illustrates another embodiment of the invention, where a singletransconductance amplifier 30 is used to provide a symmetric orasymmetric nonlinear gm, where tapped voltages selectively enablepull-up and pull-down drive stages 40-47 depending on the current outputof the transconductance amplifier 30. The drive stages 40-47 do not useany quiescent current until enabled, so the quiescent current is lowduring low gm operation. Each drive stage 40-47 may comprise a voltagedetector, for detecting an enabling voltage, and a current source. Thecurrent source may be a MOSFET connected between the load 34 and thepositive or negative rail voltage. The drive stages 40-47 are identifiedas Drive 10, 100, 1 k, and 10 k, meaning that the drive stage has apull-up or pull-down capability of 10, 100, 1 k, or 10 k times that ofthe pull-up or pull-down capability of the amplifier 30. This capabilitymay directly correlate to the relative sizes of MOSFETs used as thepull-up or pull-down devices. These scale factors are arbitrary and maybe customized for a particular application.

A series of resistors 50-53 conduct the current I1 from thetransconductance amplifier 30 to the load 34, so there are differentvoltage drops at the nodes of the resistors. These voltages are tappedby the drive stages 40-47 and are used to enable different combinationsof the drive stages 40-47 as the current I1 ramps positively ornegatively. The different drive stages 40-47 have a pair of enable pinscoupled to the taps and are enabled at different current levels outputby the transconductance amplifier 30. One enable pin on all drive stages40-47 is connected to voltage V5 as the reference voltage. The otherenable pin is connected to one of the taps. The resistor values may bethe same or different.

In the example, the currents delivered by the drive stages 40-47 areexponentially scaled, but the drive stages 40-47 can be scaled in anymanner.

In operation, at a very low current, no tapped voltage is sufficient toenable any drive stage 40-47, so the gm is only that of thetransconductance amplifier 30. Upon a positive current reaching a firstthreshold level, where the voltage drop between V1 and V5 equals theenable voltage (e.g., 1V) of the pull-up drive stage 40, the drive stage40 adds a fixed positive current to the current into the load 34. As thecurrent from the transconductance amplifier 30 is increased, the voltagedrop between V2 and V5 equals the enable voltage (e.g., 1V) of thepull-up drive stage 41, and the drive stage 41 adds its fixed positivecurrent to the current into the load 34. The process continues as thecurrent from the transconductance amplifier 30 is further increased. Asseen the positive current is exponentially (nonlinearly) increased asVin is increased due to the additional drive stages being successivelyenabled.

For negative input voltages Vin, the enable pin connections of thepull-down drive stages 44-47 are reversed so that the reference voltageV5 becomes positive relative to the other taps. The pull-down drivestages 44-47 are successively enabled as the current output by thetransconductance amplifier 30 becomes more and more negative. Thecurrent supplied to the load 34 is exponential and symmetric. The gm maybe asymmetric by selecting different drive stages. Any number of drivestages can be used to smooth out the transitions.

Since the drive stages 40-47 do not draw current before they areenabled, they do not add to the quiescent current.

Each drive stage 40-47 may include a selectable offset for itsenablement to further customize the I vs. V waveform.

If the circuit of FIG. 6 was used in a feedback loop, at steady stateconditions, only the transconductance amplifier 30 gm is relevant, socompensation for stability is easily implemented.

Further, since only one transconductance amplifier is connected to theinput voltage, the input voltage source sees a low input capacitance, soreaction time is rapid.

The added drive stages 40-47 can slew a capacitive load (including alarge MOSFET) very quickly, especially if they are exponentiallyweighted.

With capacitive loads, and where the current into the load is detectedand controlled by a feedback loop, the turning on of the various drivestages 40-47 is not directly controlled by the input voltage Vin, but bythe current slew rate through the series resistors.

FIG. 7 illustrates a circuit similar to FIG. 6, showing MOSFETs M1-M8 asthe pull-up and pull-down current sources, and a driver A1-A8 for eachMOSFET coupled to the resistor taps. Resistors R1-R4 are shown in seriesbetween the transconductance amplifier G1 and the output terminal OUT,for coupling to a load (not shown). Each driver A1-A8 has an optionalvoltage source offset V1-V8 that is used to further control when eachMOSFET M1-M8 turns on as the current generated by the transconductanceamplifier G1 ramps up or down. MOSFETs M1-M4 are P-channel MOSFETs andMOSFETs M5-M8 are N-channel MOSFETs. Varying the voltage source offsetsenables an asymmetric gm. The relative sizes of the MOSFETs M1-M8 areshown so that the variations in current supplied to the load areexponential.

FIG. 8 illustrates a circuit that may be used for each pull-up driverA1-A4 and its voltage offset. The particular configuration is for thedriver A1 having inputs connected between resistors R1 and R4. TheMOSFET M9 senses the differential voltage across resistors R1-R4. Whenthe voltage reaches the threshold voltage of the MOSFET M9, currentflows through resistor R5 to create a sufficient voltage drop to pulldown the input of the inverter U1, which pulls up on the input to theinverter U2, which then pulls down on the input of the P-channel MOSFETM1 to turn it on. In other words, when enough differential input voltageis applied to the transconductance amplifier G1 such that the currentflowing through resistors R1-R4 drops a threshold voltage (Vt), thenMOSFET M9 turns on and triggers inverters U1 and U2 to turn on MOSFETM1.A capacitor can be added across the resistor R5 to provide noisefiltering or delay. Additional inverters may be used to increase thedrive strength.

The circuit has relatively low capacitance since MOSFET M9 can be small.No quiescent current is drawn until the threshold is reached.

The pull-down drivers A4-A8 may simply be “upside down” versions of thepull-up driver of FIG. 8, with the types of MOSFET being oppositepolarity.

FIG. 9 illustrates one of the pull-down drivers A4-A8 with additionalfeatures, and the particular configuration is for the driver A8. Theadditional inverters U3-U8 increase the drive current for the largerMOSFET M8. The input senses the voltage across only resistor R4. Thecapacitor C1 across the resistor R6 serves as a noise filter or toprovide delay. The current mirror MOSFETs M11 and M12 show howlevel-shifting or analog gm can be added. By changing the size ratios ofthe MOSFETs M11 and M12, the sensitivity can be adjusted. The circuit ofFIG. 9 presents very low capacitive loading to the transconductanceamplifier G1 and consumes no quiescent current until the threshold ofMOSFET M10 is reached. All the various pull-up and pull-down circuitsmay be generally referred to as drive circuits, since they may consistof simple bipolar transistors (FIGS. 4 and 5) or may include voltagedetectors or other enablement circuits in addition to the pull-up orpull-down current sources.

All the circuits shown may be formed as integrated circuits.

Many other circuit configurations may be used to carry out the inventivetechniques.

While particular embodiments of the present invention have been shownand described, it will be obvious to those skilled in the art thatchanges and modifications may be made without departing from thisinvention in its broader aspects and, therefore, the appended claims areto encompass within their scope all such changes and modifications thatare within the true spirit and scope of this invention.

What is claimed is:
 1. A transconductance amplifier circuit forgenerating a nonlinear current comprising: a first amplifier havinginput terminals for receiving an input voltage, the first amplifierconfigured to output a first current to a load, the first current havinga substantially linear relationship to the input voltage within a firstrange so as to have a substantially linear transconductance (gm) overthe first range; at least a first resistance connected in series betweenthe first amplifier and the load, wherein the first current generated bythe first amplifier causes a voltage drop across the first resistor; anda first drive circuit having terminals coupled across the firstresistance, wherein the first drive circuit is configured to add asecond current to the first current, to supply a first composite currentto the load, when the voltage drop across the first resistance exceeds afirst threshold, such that a transconductance with respect to the inputvoltage and the first composite current is nonlinear.
 2. The circuit ofclaim 1 further comprising a second drive circuit having terminalscoupled across the first resistance, wherein the second drive circuitadds a third current to the first current, to supply a second compositecurrent to the load, when the voltage drop across the first resistanceexceeds a second threshold, such that the transconductance with respectto the input voltage and the second composite current is nonlinear. 3.The circuit of claim 2 wherein the first threshold is exceeded when apositive current through the first resistance exceeds a first level, andwherein the second threshold is exceeded when a negative current throughthe first resistance exceeds a second level.
 4. The circuit of claim 2wherein the first drive circuit comprises a bipolar NPN transistorhaving its base coupled to a first end of the first resistance and itsemitter coupled to a second end of the first resistance, and wherein thesecond drive circuit comprises a bipolar PNP transistor having its basecoupled to the first end of the first resistance and its emitter coupledto the second end of the first resistance.
 5. The circuit of claim 1wherein the first drive circuit comprises a bipolar NPN transistorhaving its base coupled to a first end of the first resistance and itsemitter coupled to a second end of the first resistance.
 6. The circuitof claim 1 wherein the first resistance comprises one of a plurality ofresistors in series between the first amplifier and the load, whereinnodes between the resistors provide different voltage taps as the firstcurrent is conducted by the plurality of resistors, the circuit furthercomprising: a plurality of first additional drive circuits havingterminals coupled to the different voltage taps, wherein differentcombinations of the first drive circuit and the first additional drivecircuits are enabled as the voltages at the voltage taps are increasedas the first current goes more positive.
 7. The circuit of claim 6wherein the terminals of the plurality of first additional drivecircuits and the terminals of the first drive circuit are enableterminals, wherein each of the drive circuits are enabled when a voltageacross its enable terminals exceeds an enable threshold.
 8. The circuitof claim 6 wherein currents supplied by the first drive circuit and thefirst additional drive circuits are weighted.
 9. The circuit of claim 8wherein currents supplied by the first drive circuit and the firstadditional drive circuits are exponentially weighted.
 10. The circuit ofclaim 6 further comprising: a plurality of second additional drivecircuits having terminals coupled to the different voltage taps, whereindifferent combinations of the second additional drive circuits areenabled as the voltages at the voltage taps are decreased as the firstcurrent goes more negative.
 11. The circuit of claim 10 wherein each ofthe first additional drive circuits comprises a first driver havinginput terminals coupled to two of the voltage taps, and wherein anoutput of the first driver drives a pull-up transistor, and wherein eachof the second additional drive circuits comprises a second driver havinginput terminals coupled to two of the voltage taps, and wherein anoutput of the second driver drives a pull-down transistor.
 12. Thecircuit of claim 6 wherein each of the first additional drive circuitscomprises a first driver having input terminals coupled to two of thevoltage taps, and wherein an output of the first driver drives a pull-uptransistor.
 13. The circuit of claim 12 wherein a voltage offset iscoupled to one of the input terminals of the first driver.
 14. Thecircuit of claim 12 wherein the transconductance with respect to theinput voltage and the first composite current is nonlinear andasymmetrical.
 15. The circuit of claim 6 wherein the first additionaldrive circuits are enabled sequentially as the first current isincreased.
 16. The circuit of claim 1 further comprising a voltageoffset coupled to one of the terminals of the first drive circuit. 17.The circuit of claim 1 wherein the first drive circuit consumes nocurrent until the first drive circuit generates the second current. 18.A method performed by a transconductance amplifier circuit generating anonlinear current comprising: generating a first current by a firstamplifier having input terminals for receiving an input voltage, thefirst current having a substantially linear relationship to the inputvoltage within a first range so as to have a substantially lineartransconductance (gm) over the first range; dropping a voltage across atleast a first resistance connected in series between the first amplifierand a load, due to the first current flowing through the firstresistance; and adding a second current to the first current, by a firstdrive circuit having terminals coupled across the first resistance, tosupply a first composite current to the load, when the voltage dropacross the first resistance exceeds a first threshold, such that atransconductance with respect to the input voltage and the firstcomposite current is nonlinear.
 19. The method of claim 18 furthercomprising: adding a third current to the first current by a seconddrive circuit having terminals coupled across the first resistance, tosupply a second composite current to the load, when the voltage dropacross the first resistance exceeds a second threshold, such that thetransconductance with respect to the input voltage and the secondcomposite current is nonlinear.
 20. The method of claim 19 wherein thefirst threshold is exceeded when a positive current through the firstresistance exceeds a first level, and wherein the second threshold isexceeded when a negative current through the first resistance exceeds asecond level.